Scalable, multi-layer MIMO transceiver

ABSTRACT

Disclosed herein is an innovative multi-layer hybrid/digital MIMO architecture that comprises single-stream or fully-connected (FC) multi-stream beamforming tiles (with RF complex-weights) in the first layer, followed by a fully connected (analog/digital) baseband layer. This architecture overcomes the complexity versus spectral-efficiency tradeoffs of existing hybrid MIMO architectures and enables MIMO stream/user scalability, superior energy-efficiency, and spatial-processing flexibility.

RELATED APPLICATIONS

This application claims the benefit of U.S. Provisional PatentApplication Ser. No. 62/967,616, filed Jan. 30, 2020, the contents ofwhich are incorporated herein in their entirety. In addition, thisapplication is a continuation-in-part of U.S. patent application Ser.No. 17/113,288, filed Dec. 7, 2020, the contents of which areincorporated herein in their entirety.

GOVERNMENT RIGHTS

This invention was made with U.S. government support under CNS1823235awarded by the National Science Foundation. The U.S. government hascertain rights in the invention.

BACKGROUND OF THE INVENTION

Multi-input-multi-output (MIMO) communication, along with beamforming atmillimeter-wave frequencies, is a key element in beyond-5G wirelesssystems to simultaneously improve data-rate and channel capacity.Prototypes are currently being developed that support a wide variety ofarchitectures starting with basic single-stream, phased-arraybeamforming at a single mm-wave frequency band, to multi-antennapolarization MIMO, multi-stream MIMO at a single frequency band, andreconfigurable multi-stream MIMO and multi-antenna inter-band carrieraggregation (CA) across two frequency bands. To realize multi-streamoperation while performing energy-efficient RF domain spatial signalprocessing, two kinds of hybrid beamforming architectures are beingconsidered of the partially connected (PC) and fully connected (FC)types. While PC structures are scalable, FC achieves superiorspectral-efficiency, transmit and receive mode energy efficiency, andcarrier aggregation (CA) from the entire aperture. However, scaling suchFC architectures for more than two streams imposes significantchallenges.

SUMMARY OF THE INVENTION

Disclosed herein is a novel multi-layer hybrid beamforming MIMOarchitecture that comprises multiple tiles of single-stream partiallyconnected (PC-tile) or multi-stream fully connected (FC-tile)beamformers with RF-domain complex-weighting in a first layer, followedby one or more fully-connected additional layers in the analog/digitalbaseband domain. This architecture overcomes the complexity versusspectral-efficiency tradeoffs between single-layer PC and FC hybrid MIMOarchitectures and enables efficient upward scaling of the number ofsupported antennas as well as streams.

Further, low-complexity RF-tiles with additional baseband processingallows this architecture to achieve excellent energy-efficiency andalgorithmic flexibility. It is also important to note that the use ofFC-tiles instead of PC-tiles in the multi-layer architecture confersmultiple additional advantages, such as improving the spectralefficiency, supporting CA from the full antenna aperture and full-duplexbeamforming with per-element self-interference cancellation. Thearchitectural tradeoffs, design concepts, and system simulations arealso discussed herein.

Another key feature of the disclosed invention is a compact transceiverarchitecture having innovative circuit techniques that enable highlyreconfigurable bidirectional operation. To this end, a Cartesiansplitting based transmit beamforming architecture is first introduced,and then merged with the Cartesian combing based receive beamforming forcompactness and dual-band operation. Further, the transceiver circuitarchitecture features a bidirectional beamforming network with passivestructure reuse, bidirectional dual-band frequency translationcapability, reconfigurable dual-band LO generation and distribution,Cartesian analog-domain second-layer beamforming, and a newphase-invariant constant-current programmable gain amplifier design. Todemonstrate the aforementioned systemic, architectural, and circuitconcepts, the design of a 28/37 GHz eight-element two-tile four-streamtransceiver prototype is shown.

The invention has applicability in communication, radar and imagingsystems, including, for example, 5G and beyond-5G wireless networks,fixed-wireless access networks, short-reach wireless access networks,wireless backhaul networks, Wi-Fi wireless networks, automotiveradar/imaging, V2X communication/radar, autonomous transportationsystems, satellite communication and many others.

BRIEF DESCRIPTION OF THE DRAWINGS

The patent or application file contains at least one drawing executed incolor. Copies of this patent or patent application publication withcolor drawing(s) will be provided by the Office upon request and paymentof the necessary fee.

FIG. 1A shows signal-flow graph representation of various MIMObeamforming transceiver architectures, including digital beamforming,and single- and multi-layer hybrid beamforming. The architectures arecompared in terms of spectral efficiency, scalability, complexity,beamforming flexibility, and energy efficiency. FIG. 1B shows threevariations of the primary embodiment of the invention.

FIG. 2 is a schematic of a multi-layer HBF concept and an example for aneight-element two-stream architecture.

FIG. 3A is a schematic of a vector-modulator-based Cartesian phaseshifting single-element transmitter. FIG. 3B is a schematic of amodified vector modulator architecture wherein RF splitting inside thequadrature splitter is translated to baseband. FIG. 3C is a schematic ofa Cartesian-splitting-based beamforming transmitter architecture whereina 90° phase shifter is absorbed in the LO path from RF. FIG. 3D is aschematic of a Cartesian-splitting-based phase shifting for one elementand two baseband streams wherein one stream is shown in blue and theother steam is shown in red.

FIG. 4 is a simplified schematic of two-layer, eight-element,four-stream hybrid beamforming transceiver prototype in accordance withthe present invention.

FIG. 5 is a schematic of dual-band, bi-directional front-end consistingLNA, PA and antenna interface switch.

FIG. 6A is a schematic of a reconfigurable bidirectionalsplitting/combining network.

FIG. 6B is a schematic of a programmable bidirectional transconductor.

FIG. 7A is a schematic of a simplified lumped element model; FIG. 7B isa schematic of a differential half circuit equivalent representation.FIG. 7C is a graph showing effective coupling factor (k_(E)) andself-inductance (L_(E)) across transformer's coupling factor (k_(X)).FIG. 7D is a graph showing transimpedance (Z₂₁) amplitude response fork_(X)=0.6 and k_(X)=0.3.

FIG. 8A is a schematic for a bi-directional complex quadrature mixer;FIG. 8B is a schematic for an up-conversion mixing core w/o the loadinductor; and FIG. 8C is a schematic of a quadrature downconversionmixer.

FIG. 9 is a dual-band quadrature LO generation and distribution networkfor the two-tile four-stream MIMO HBF transceiver shown in FIG. 4 .

FIG. 10A is a schematic of a baseband cartesian complex weighting andcombining;

FIG. 10B is a schematic of a programmable sign-switchable transconductorthat maintains the input and output impedance and quiescent currentconsumption across the tuning range.

FIG. 11A is a schematic of a STAR system for multiple TX and RXantennas. FIG. 11B is a schematic of a STAR system for phased arrayantennas.

FIG. 12 is a schematic of a simultaneous transmit and receivebeamforming architecture. The example shown is for a 16-element antennaarray consisting four FC-tiles.

FIG. 13 is a schematic of a STAR beamforming system consisting N TXantennas and N RX antennas with three-step successive self-interferencecancellation (SIC) mechanism: the RF-domain per-element SIC before RXbeamforming, null-steering based SIC in the RX beamforming, and RF-SICafter RX beamforming.

FIG. 14 is a schematic for a full duplex prototype with three-stepcancellation shown for four elements. Circuit architecture isreconfigurable between half-duplex MIMO mode and full-duplex mode.

FIG. 15 is a schematic for an LMS-based RF-domain per-element SIC weightadaptation with time-multiplexed error extraction. SIC adaptation isshown for four RX elements.

FIG. 16A is a schematic for the digital adaptation circuitry, shown fora single element scenario; FIG. 16B is a schematic for the basebandequivalent representation of SIC weight adaption loop.

FIG. 17 is a schematic showing Step #1 or Step #3SIC with digital domaingroup delay correction.

DETAILED DESCRIPTION

MIMO systems allow the transmission or reception of multiplesimultaneous data streams by means of independent beamforming for eachstream. This section reviews existing MIMO system architectures anddescribes the proposed multi-layer architecture. Starting withsignal-flow graph representations, as shown in FIG. 1A, thearchitectures are then compared in terms of spectral efficiency,scalability, complexity, beamforming flexibility, and energy efficiency.

Digital and Single-Layer Hybrid Beamformers

In digital beamformers (DBFs), spatio-temporal or spatio-spectral signalprocessing for N_(S) data streams is performed by a digital signalprocessor (DSP) which is interfaced to an N_(A)-element antenna array byhaving one frequency translation chain and analog-digital conversioninterface per element (i.e., N_(RF)=N_(A)). DBFs offer completeflexibility in terms of the number of supported streams and their beampatterns while also achieving the best spectral efficiency. Moreover,since the RF, analog-baseband and analog-digital interfaces in DBF's aremodular, DBFs can in principle be easily scaled to large N_(A), giventhe availability of a sufficiently capable DSP. However, DBF's sufferfrom high power and chip area for consumption in large antenna arrays,not just in the RF/analog portions, but also in the DSP for highthroughput scenarios. Moreover, in DBF receivers, since no spatialfiltering occurs before digitization the dynamic range and theeffective-number-of-bits specifications of the ADC increase drastically.

Hybrid beamformers (HBF), on the other hand, perform extensive spatialsignal processing in the RF domain, and have relatively few frequencytranslation chains and analog-digital interfaces to support digital MIMOprocessing. The number of frequency translation chains N_(RF) is greaterthan but commensurate with N_(S) which in turn is typically much smallerthan N_(A), (i.e., N_(A)>>N_(RF)≈>N_(S)). Therefore, HBFs offer betterarea and energy efficiency than DBFs for a given N_(S). FIG. 1Aillustrates several HBF architectures using a pseudo signal-flow graphrepresentation. Architectures H1-H3 feature a single spatial signalprocessing layer. The single-layer partially-connected (PC) architectureH1 is equivalent to having multiple phased array/RF-beamformer tiles(K=# tiles), one for each MIMO stream (hence, K=N_(RF)=N_(S)). Althoughthis architecture has low complexity and can easily be scaled tomultiple antennas/streams due to its modular tiled approach, in H1, onlya subset of antenna elements is connected to each stream, therebyresulting in reduced per-stream beamforming gain. On the other hand, inthe fully-connected (FC) architecture H2, each baseband stream isconnected to the full aperture. Hence, H2 can achieve the same spectralefficiency as a DBF for identical N_(S). Scaling H2 to support a greaternumber of streams is challenging. Therefore, the H1 and H2 architecturefaces a spectral-efficiency versus scalability trade-off. This trade-offcan be relaxed to a certain extent by using a single-layer HBFarchitecture with FC tiles (H3 in FIG. 1A, referred to as the hybridconnected structure). Here, K low-complexity FC tiles, each supportingtwo streams, are used to support N_(S)=2K MIMO streams. For each stream,the H3 architecture achieves twice the beamforming gain compared to H1.However, there is still a significant performance gap compared to H2 orDBF. Additionally, it is important to note that, single-layer HBFs(H1-H3) RF-domain-only spatial signal processing offer limitedflexibility to implement beam acquisition, training and trackingalgorithms compared to DBFs.

Multi-Layer Hybrid MIMO Architecture

The Multi-Layer Hybrid MIMO Architecture of the present invention uses KPC or FC tiles (H4 in FIG. 1A and H5, respectively in FIGS. 1A and 1B)in the first layer which interfaces N_(A) antennas with N_(RF) frequencytranslation chains (N_(RF)=K for H4; N_(RF)=2K for H5 with two-stream FCRF tiles). A second layer implements fully-connected beamforming fromN_(RF) baseband signals to N_(S) streams at RF or baseband, in analog ordigital domain. In contrast to the single-layer FC-HBF architectures,the new multi-layer architectures retain low RF-domain signal processingcomplexity while enabling upward scaling of N_(S). Its spectralefficiency is superior to single-layer PC-HBF and approaches that ofDBF.

FIG. 1B shows three variants of the primary embodiment of the invention,the first variant is a two-layer hybrid architecture in which thebaseband processing in the second layer occurs in the analog domain. Thesecond variant is a two-layer architecture in which the basebandprocessing occurs in the second layer in the digital domain. The thirdembodiment is a generalized three-layer architecture in which thebaseband processing occurs in a second, analog domain layer and a third,digital domain layer.

The two-layer MIMO operation can be understood intuitively from asimplified case of PC-tile two-stream RX architecture with N_(A)=8 andK=N_(RF)=N_(S)=2, as shown in FIG. 2 . The spectra of two differentincoming data streams from two separate angles of arrival are indicatedin red and blue at various points in the signal path. The weights ineach PC tile are set to synthesize a single composite array pattern thathas two “main” lobes (N_(S) main lobes in general for N_(S) stream),each directed towards one incoming stream. Example beam patterns areshown in the bottom portion of FIG. 2 . Hence, in the first layer, alltiles perform SNR improvement for all incoming streams, all tilesspatially filter out signals that have an angle of arrival other thanall the streams and all tiles perform no cross-stream isolation.However, although all tiles receive all the streams, as many such tilesare present in the first layer, some spatial information remains thatcan be further utilized. Therefore, in the fully-connectedanalog/digital baseband second layer, the weights can be set to achievespatial separation of streams, and hence, cross-stream cancellation(example beam patterns are shown) and further improvement of SNR byappropriately adding signals from multiple tiles. Similarly, amulti-layer PC-tile TX array operation can be understood as thefollowing. Each tile sends out signals simultaneously in many differentdirections for many streams. Now, the input to each tile from basebandis already pre-processed in such a manner that each stream aftercumulative effect from all tiles constructively adds in only one of themany transmit directions.

Multi-layer HBF architectures are not subject to the trade-off betweendifferent single-layer HBFs and offer the following advantages: (1)superior energy-efficiency as compared to DBF due to RF heavy processingand only a limited number of frequency translation stage; (2) lowerdesign complexity than the H2 architecture due to the use of RF-tileswith only one/two streams; (3) better scalability than the H2architecture both in case of N_(A) and N_(S) due to the modular tiledapproach of RF processing; (4) superior spectral efficiency as comparedto the H1 architecture; and (5) better design flexibility than allsingle-layer HBF architectures because of the second layer of basebandprocessing.

The use of low-complexity FC tiles instead of PC tiles in a multi-layerHBF architecture confers numerous advantages as follows: (1) As shown inthe insert in FIG. 2 , an FC-tile multi-layer HBF can support a highernumber of streams compared to a PC tile, with each stream having thesame performance as in the case of the PC tile. (2) Each stream in theFC tile can support different channels or bands, thereby enablingcarrier-aggregation (CA) from the full aperture while also performingMIMO at each carrier. Full-aperture CA cannot be supported in the caseof PC-tile. (3) The multi-layer HBF with FC tile enables front-endself-interference cancellation (SIC) that cancels the TX leakage in theRX path at RF on a per-element basis. Such SIC cannot be supported in aPC-tile.

Transceiver Architecture

Cartesian-Combining MIMO/Beamforming Receive Path

The receive path employs the Cartesian-combining architecture, which wasfirst proposed for single-stream RF beamforming with homodynedown-conversion, and later extended to hybrid MIMO reception. Aheterodyne embodiment having image filtering, but no image cancellationwas demonstrated. A generalized heterodyne embodiment withreconfigurable image cancellation was also demonstrated; using a singleLO generation sub-circuit in each downconversion chain that supportedMIMO reception at either 28 GHz or 39 GHz, or concurrently in bothbands.

In the present invention, the receive signal path is designed to supportboth the 28 and 39 GHz bands, either solely or concurrently. However, adirect conversion approach is adopted, thereby requiring a dedicated LOdistribution circuit for each downconversion chain but avoiding the needfor image-reject calibration.

Cartesian-Splitting MIMO/Beamforming Transmit Path

The Cartesian-splitting beamforming transmitter architecture will now bedisclosed. FIG. 3A shows a quadrature upconverter followed by a vectormodulator complex-weighting circuit and a power amplifier driving anantenna element. One goal of the present invention is to eliminate thequadrature splitter, typically implemented as a quadrature hybrid orpolyphase filter, from the signal path. This can be done by twotransformations: first, by commutating the RF splitting and thequadrature upconversion operations, as shown in FIG. 3B, and, second, byabsorbing the 90° phase shifter in the RF signal path into the LO portsof the mixers and appropriately modifying the connections of the LOphases, as shown in FIG. 3C. The Cartesian-splitting complex-weightingprinciple can also be understood mathematically. The complex basebandsignal, {tilde over (x)}_(BB) ≡x_(BBI) (t)+jx_(BBQ)(t), is upconvertedby a complex quadrature mixer, producing a complex-valued signal withreal and imaginary parts u_(r) (t) and u_(i) (t) respectively:u(t)={tilde over (x)} _(BB)(t)e ^(2πf) ^(LO) ^(t) ≐u _(r)(t)+ju _(i)(t)where:u _(r)(t)=x _(BBI)(t)C−x _(BBQ)(t)Su _(i)(t)=x _(BBI)(t)S+x _(BBQ)(t)CC≡cos(2πf _(LO) _(t) )S≡sin(2πf _(LO) _(t) )   (1)

In a phased array or partially-connected MIMO transmitter, each antennamust be driven by an independently complex-weighted signal. Thus, thesignal driving the k^(th) antenna (or the k^(th) PA) can be generated byindependently gain-scaling the real and imaginary parts and summing themtogether, as shown in Eq. (2a) below.(2a):x _(k)(t)=A _(r) u _(r)(t)−A _(i) u _(i)(t)=Re[(A _(r) +jA_(i))u(t)](2b):=Re[(A _(r) +jA _(i)){tilde over (x)} _(BB)(t)e ^(2πf) ^(LO) ^(t)]=A _(r) {x _(BBI)(t)C−x _(BBQ)(t)S}−A _(i) {x _(BBI)(t)S+x _(BBQ)(t)C}  (2)

Equivalently, Eq. (2b) shows that the envelope of the bandpass signalx_(k)(t) is equal to the complex baseband envelope scaled by acomplex-valued weight (A_(r)+jA_(i)). In practice, gain scaling of thereal and imaginary parts can be implemented by a pair of programmablegain transconductors, while the summing can be implemented by combiningtheir output currents.

It is straightforward to extend the aforementioned principle to afully-connected hybrid MIMO transmitter. As shown in FIG. 3D, for asingle antenna in a two-stream FC-MIMO transmitter, a separateupconversion chain is required for each stream. To apply a complexweight for the s^(th) stream to the k^(th) antenna, the real andimaginary outputs of the s^(th) chain are scaled with a pair of PGAsA_(r;k,s) and A_(i;k,s). The weighted streams are then summed togetherbefore the k^(th) antenna. In the transmitter of to present invention,summing is done at the input of the k^(th) PA; this signal can bewritten as:

$\begin{matrix}{{x_{k}(t)} = {{Re}\left\lbrack {\sum\limits_{s = 1}^{S}{\left\{ {A_{{r;k},s} + {jA_{{i;k},s}}} \right\}{{\overset{˜}{x}}_{{BB},s}(t)}e^{2\pi f_{LO^{t}}}}} \right\rbrack}} & (3)\end{matrix}$Bi-Directional MIMO Transceiver

In the present invention, the Cartesian-combining receive path andCartesian-splitting transmit path are combined to implement a hybridMIMO/beamforming transceiver, which is shown schematically in FIG. 4 .This circuit architecture has several advantages. First, it avoids theuse of RF-domain phase shifters or quadrature hybrids that are oftenbulky, lossy, and narrowband. For each transmitter stream, the basebandcircuitry and the upconversion chain is shared between all antennaelements. Thus, the overhead incurred in a Cartesian-splitting MIMOtransmitter (relative to a single antenna transmitter) is a pair ofPGAs. Second, high resolution, calibration-free, digitally programmableCartesian complex-weighting with low gain and phase error can beimplemented. Third, and most importantly, the weighting principle isinherently wideband, and can be realized at any frequency where thefront-end PGAs achieve sufficient gain. Therefore, this architecture iswell suited for wideband application, and also for dual-band applicationwith widely separated frequencies.

Transceiver Prototype Design

A 28/37 GHz two-layer hybrid beamforming MIMO transceiver prototype witheight elements and four chains has been designed to demonstrate theconcepts described herein. The prototype is of type H5, as shown in FIG.1A and FIG. 1B, with parameters N_(A)=8, K=2, N_(RF)=4, and N_(S)=4. Asimplified schematic is shown in FIG. 4 . The first layer comprises twofour-element, two-stream fully-connected tiles. In each tile, theantenna port is interfaced with two frequency conversion chains viafront-end amplifiers and PGA's for complex-weighting. In the secondlayer, bi-directional spatial signal processing is performed at analogbaseband. In the receive path, the four downconversion chains connect toCartesian complex weights whose outputs represent the received streams(or a mixture of the received streams, with one stream dominant). In thetransmit path, four analog baseband data streams are weighted,upconverted using four upconversion chains and applied to the firstlayer tiles

The Cartesian-combing/splitting technique described in the previoussection is used to perform RF-domain first-layer beamforming in eachstream of the FC tile (the two streams in each tile are shown in red andblue in FIG. 4 ). In addition to its architectural advantages, theCartesian-combining/splitting technique is well-suited to dual-bandbeamforming. This is because the RF-domain network is only based onprogrammable transconductors and current-mode combiners or voltage-modesplitters and can all be designed to have dual-band frequency response.The frequency translation chains in each tile can select either 28 or 37GHz LO, thereby providing maximum re-configurability betweencarrier-aggregation and MIMO modes. Compared to the heterodyneimage-reject dual-band beamforming architecture, which minimizes LOtuning range, the homodyne architecture is used here that is simpler,offers more flexibility in LO frequency selection and bettercross-stream isolation at the expense of larger LO tuning requirement.

The prototype multi-layer HBF transceiver comprises the followingblocks: (1) one LNA/PA dual-band bidirectional interface per elementshared between the two streams in each tile, (2) dual-band bidirectionalbeamforming network with shared passives between the TX and RX path forcompactness, (3) one homodyne complex-quadrature up/down conversionstage per stream, (4) dual-band LO generation and distribution network,(5) baseband TX and RX second layer analog domain processing, and (6)digital control and adaptation circuitry. Design considerations of thekey blocks are described next.

LNA and Bi-Directional Antenna Interface Design

Each RF signal port of the chip connects to a dual-band, bidirectionalfront-end circuit which interfaces an antenna element to the firstbeamforming layer, as shown in FIG. 4 . The front-end, shown in FIG. 5 ,comprises a low-noise amplifier, a power amplifier, and an antennainterface network which combines a T/R switch function, a powercombining function in transmit mode and an input matching function inthe receive mode. The fully-connected first layer achieves better energyefficiency than a partially-connected type when PA's with betterback-off efficiency than Class-A are used. Therefore, the PAs in thepresent front-end employ a two-way power combining topology with Class-Bunit PA's in the output stage, each having a second harmonic shortingnetwork; this enables better peak and back-off efficiency. The front-endcan be switched between transmit and receive modes by means of threeswitches, as shown in FIG. 5 . The output side series power combiningnetwork also constitutes input matching network to the LNA which uses aG_(m)-boosted common-gate input stage. All gain stages except the inputLNA stage and the PA input stage use common-source differential pairswithout a tail current source but with cross-coupled capacitiveneutralization for improved differential mode stability. These stagesuse transformer coupled-resonator loads to obtain dual-band frequencyresponses. To equalize gains in the two bands, one gain stage in each ofthe LNA/PA paths uses driving port impedance (Z₁₁) load; all otherstages use trans-impedance (Z₂₁) loads. In this prototype, the gain ofthe LNA is increased by adding an additional gain compensation stagethat improves the overall gain by 4 dB in 28 GHz band and 11 dB in 37GHz band.

First Beamforming Layer Network

The RF-domain beamforming network in the first layer is constructed bycombining bi-directional sections similar to FIG. 6A, following thesystem shown in FIG. 4 . The section shown in FIG. 6A comprises threebidirectional transconductor stages (G1-G3) connected to a coupledresonator. Transconductors G1-G3 use cross-connected differential pairsone of which is turned ON for forward or reverse signal flow (in thecomplex-weighting sections, these are binary-weighted, digitallyswitched). Thus, the signal path can be reversed without having switchesin the RF signal path, thereby improving losses and bandwidth. Thesecondary side of the coupled resonator serves as a voltage-splitter inreceive mode (with G1-G3 in forward mode) and a current-combiner load intransmit mode (with G1-G3 in reverse mode). Using this structure in lieuof the traditional Wilkinson structure provides another advantage,namely that a third mode is available for current-mode self-interferencecancellation for simultaneous transmit-receive operation where G1 and G2are set in forward mode and G3 in reverse mode. Passive structuresthroughout the first-layer network are shared between the TX and RXsignal paths in order to reduce die area. The resulting compactness alsohelps minimize interconnect losses, and hence overall power consumption.

Front-End and First Layer Frequency Response

The front-end and the first-layer beamforming network is designed tosupport concurrent operation in the 28 and 37 GHz bands. Their frequencyresponses are tailored by adjusting the coupling coefficient (and hencethe poles) of the coupled resonators. A moderate coupling factor(k_(X)=0.25-0.35) is chosen for the coupled-resonators in the LNA/PAstages. In the beamforming network, however, the presence of relativelylong transmission lines (100-300 um) between the active circuits and thecoupled resonators loads lead to somewhat different designconsiderations, as discussed next. Specifically, a substantially highercoupling coefficient (k_(X)=0.4-0.6) is required to achieve therequisite dual-band frequency response. The design approach starts withanalysis using lossless lumped-element models for the transmission lineinterconnect (parameters L_(L), k_(L), C_(L), and C_(M)) and thetransformer (parameters L_(X) and k_(X)). The circuit is shownschematically in FIG. 7A, and its differential-mode half-circuitequivalent is shown in FIG. 7B. The voltages V₁, V₂ and currents I₁, I₂in FIG. 7B are related as follows:

$\begin{matrix}{{V_{1} = {{{S\left\lbrack {L + {\left( \frac{L_{X}}{2} \right)\left( \frac{F_{1}}{F_{2}} \right)}} \right\rbrack}I_{1}} + {{S\left\lbrack {{k_{x}\left( \frac{L_{X}}{2} \right)}\left( \frac{1}{F_{2}} \right)} \right\rbrack}I_{2}}}}{V_{2} = {{{S\left\lbrack {{k_{x}\left( \frac{L_{X}}{2} \right)}\left( \frac{1}{F_{2}} \right)} \right\rbrack}I_{1}} + {{S\left\lbrack {L + {\left( \frac{L_{X}}{2} \right)\left( \frac{F_{1}}{F_{2}} \right)}} \right\rbrack}I_{2}}}}{{{where}: F_{1}} = {1 - {{\omega^{2}\left( \frac{CL_{X}}{2} \right)} \times \left( {1 - k_{X}^{2}} \right)}}}{F_{2} = {\left\lbrack {1 - {{\omega^{2}\left( \frac{CL_{X}}{2} \right)}\left( {1 + k_{X}} \right)}} \right\rbrack\left\lbrack {1 - {{\omega^{2}\left( \frac{CL_{X}}{2} \right)}\left( {1 - k_{X}} \right)}} \right\rbrack}}} & (4)\end{matrix}$

If the circuitry inside the blue dashed rectangle in FIG. 7B isrepresented as an equivalent transformer with coupling factor of k_(E)and self-inductance of L_(E)/2, it can be derived from Eq. 4) that:

${L_{E} = {{2L} + {L_{X} \times \left( \frac{F_{1}}{F_{2}} \right)}}}{k_{E} = \frac{k_{X}}{F_{1} + {F_{2} \times \frac{2L}{L_{X}}}}}$

-   -   (5)

Therefore, in the presence of the transmission lines, the effectivetransformer coupling factor is reduced, and self-inductance isincreased. FIGS. 7 (C, D) illustrate an exemplary scenario where two 240um transmission lines are symmetrically connected to a transformer(L_(X)=100 pH and k_(X) is tunable). The transmission line interconnectis first simulated using a field solver and its lumped elementparameters are extracted as L_(L)=155 pH, k_(L)=0.52, C_(M)=15 fF, andC_(L)=40 fF. It can be seen from FIG. 7C that the simulated L_(E) andk_(E) variation with k_(X) closely matches the simplified analysis inEq. (5). As further verified in FIG. 7D, the transimpedance of thecomposite transmission line and transformer system with moderatecoupling coefficient (k_(X)=0.3) has low bandwidth since the effectivecoupling coefficient is low (k_(E)=0.17). On the other hand, the systemcan be made to operate as a dual-band load when a relatively highcoupling factor k_(X)=0.6 is chosen, which results in moderate effectivecoupling (k_(E)=0.325).

Frequency Translation Chain Design

FIGS. 8A-C show a detailed schematic of the bi-directional complexquadrature mixing stage. In each tile, the up- and down-conversionmixing paths are connected to the beamforming network by a pair ofcoupled resonators (one each for the real and imaginary paths). Theprimary port of each coupled resonator connects to the bidirectionaltransconductors in the beamforming network. The secondary port of eachcoupled resonator acts as a load of a quadrature up-conversion stage,and also as a splitting node to a quadrature downconversion stage.

The downconverted outputs of the quadrature mixer pairs in the receivepath are weighted (B₁-B₈ in FIG. 8A) and summed to generate basebandquadrature outputs. The weights B₁-B₈ serve two purposes: first, toperform the output summing operation required to completeCartesian-combining in the first beamforming layer; and second, tocorrect for gain and phase errors. The nominal B₁-B₈ values in theabsence of gain/phase errors are shown in FIG. 8A. Including gain errors(Δ₁-Δ₄) and phase errors (θ₁-θ₄) of the four down-conversion mixingpaths, the B₁-B₈ values are given below:

$\begin{matrix}{{B_{1} = \frac{{\cos\theta}_{2}}{\Delta_{1}C_{1}}}{B_{2} = \frac{- {\sin\theta}_{2}}{\Delta_{1}C_{1}}}{B_{3} = \frac{{\sin\theta}_{1}}{\Delta_{2}C_{1}}}{B_{4} = \frac{{\cos\theta}_{1}}{\Delta_{2}C_{1}}}{B_{5} = \frac{- {\cos\theta}_{4}}{\Delta_{3}C_{2}}}{B_{6} = \frac{- {\sin\theta}_{4}}{\Delta_{3}C_{2}}}{B_{7} = \frac{- {\sin\theta}_{3}}{\Delta_{4}C_{2}}}{B_{8} = \frac{{\cos\theta}_{3}}{\Delta_{4}2}}{{{where}: C_{1}} = {\cos\left( {\theta_{1} - \theta_{2}} \right)}}{C_{2} = {\cos\left( {\theta_{3} - \theta_{4}} \right)}}} & (6)\end{matrix}$

This scheme can correct for several types of gain/phase errors(θ₁-θ₄/Δ₁-Δ₄) including: (1) quadrature gain/phase error between the Iand Q LO paths, (2) gain/phase error within the two I paths or the two Qpaths of the LO, and (3) gain/phase error between the real and imaginarysignal paths.

Similar to the RX path, the A₁-A₈ weights in the TX path are applied tothe baseband I/Q data streams to perform complex mixing and gain/phaseerror correction before being input to the upconversion mixers. Thetransistor level schematic for the up-conversion mixing core (excludingthe load transformer) and the quadrature down-conversion mixer are shownin FIG. 8B and FIG. 8C, respectively. Current steering is used in boththe up- and down-conversion mixing stage to improve linearity and noiseperformance.

Dual-Band LO Generation and Distribution Network

Because this transceiver employs direct conversion in both frequencybands, the LO distribution circuitry must have a frequency response thatcovers both bands. Moreover, because is desirable for the band for eachstream (i.e., each up/downconversion pair) to be independentlyselectable, dedicated LO generation/distribution units are necessary foreach stream. In this transceiver, LO synthesis circuitry is notincluded. However, it accepts a single-ended LO signals in the 28 GHzand 39 GHz, from which differential-quadrature signals appropriate forfeeding the active mixers shown in FIG. 8B are generated anddistributed. The LO subsystem is shown in FIG. 9 . The 28 and 39 GHz LOinputs are converted to differential using baluns, then buffered and fedto coupled resonator quadrature hybrids (CRQH). In contrast to otherquadrature generation methods, the CRQH's are extremely compact, havelow loss and easily implemented in differential form. The limitedbandwidth over which the CRQH maintains 90° phase difference is not anissue here because the CQRH can be tuned with along the LO frequency. Inthis transceiver, separate CRQH's are used for each band. The voltageoutputs of the CRQH's are buffered by transconductors terminated indual-band coupled resonator loads, as shown in FIG. 9 . The LO fed to aparticular stream is selected to either the 28 or 39 GHz by switching onor off the appropriate buffer transconductor.

Second (Baseband) Beamforming Layer Design

Cartesian Baseband Beamforming: The second beamforming layer usesseparate Cartesian complex weights at baseband for the transmit andreceive paths (See FIG. 4 ). A detailed schematic of a section of thesecond beamforming layer is shown in FIG. 10A, where Cartesian complexweights are applied to two baseband streams and their outputs summedtogether, in accordance with the architecture of FIG. 4 . Each complexweight is realized using four programable transconductors whose outputcurrents are combined with appropriate polarity, as shown in FIG. 10A.In-phase and quadrature components of output currents from two suchcomplex weights are summed together and converted to voltage usingresistive loads (which use current bleeding to set the common-modelevel). Each programable transconductor has 6-bit (including sign bit)binary gain control, thereby enabling fine resolution basebandbeamforming.

New PGA Architecture: An improved topology was sought for theprogrammable transconductor due to the large number of such cells and toachieve high complex weighting accuracy over all possible settings.Desirable attributes of a programmable transconductor include preciselinear increments, constant input/output capacitances and outputresistance (to ensure phase invariance across settings), low hardwareoverhead and constant output common-mode voltage. The phase-invarianttopology has the first two attributes. However, to present constantinput capacitance, the design uses additional analog circuit blocks.Moreover, its common-mode voltage is a function of the gain setting. Toovercome these limitations, the topology shown in FIG. 10B wasdeveloped, along with a digital encoding scheme that ensured constantcapacitance, output resistance and DC current.

The proposed transconductor is a binary-weighted array ofcross-connected differential pairs. Each cell in the array can be turnedon or off by switching the bias of that cell's tail current source. Adigital controller controls how the cells are turned on or off,according to an algorithm discussed next. Suppose the desired weight isrepresented as W, and the digital controls for the positive and negativepolarity cells are DIGP and DIGN, respectively. Let the maximum value ofDIGP or DIGN be N. The controller selects a set of value for DIGP orDIGN such that:DIGP−DIGN=WDIGP+DIGN=(N) or (N+1)   (7)

That is, the total number of turned-on cells is roughly constant (withan unavoidable one LSB variation) across the entire tuning range (+N to−N). The solution to Eq. (7) can be shown to be the following:

$\begin{matrix}{{{DIGP} = \left\lfloor \frac{N + 1 + W}{2} \right\rfloor}{{DIGN} = \left\lfloor \frac{N + 1 - W}{2} \right\rfloor}} & (8)\end{matrix}$

Consider two example cases with W=31 and W=−15, where N=31. According toEq. (7), the DIGP and DIGN values for the first case is 31 and 0respectively, and, for the second case, is 8 and 23, respectively. Basedon the above design strategy, as the total turned-on cells only vary byone LSB, the input capacitance, output impedance, and DC currentvariation are also bounded to one LSB variation between the on and theoff cell. The digital control circuitry that implements thefunctionality in Eq. (8) is shown in FIG. 10B.

System Considerations for Full-Duplex (FD) Beamforming (BF) Operation

Simultaneous transmit-and-receive (STAR) systems can be classified intothe frequency-division duplex (FDD) type or the full-duplex (FD) type.In both classes, leakage of the large transmit signal can severelycorrupt the weak received signal. Suppression of self-interference (SI)can be achieved by a combination of signal isolation and signalcancellation. In the FDD case, some isolation is naturally availablebecause the transmit and receive signals occupy different frequencybands. This mechanism is not available in FD systems, which makes the FDcase significantly more challenging. In both cases, isolation is mostsimply achieved by using separate antenna transmit and receive arrays.SI cancellation, on the other hand, can be implemented by introducingweighted and/or filtered replicas of the transmit signal to successivelycancel the SI at various points along the receive path.

SIC Dimensionality Reduction for Phased Array System

In a multi-antenna system with N_(S) streams and N TX and RX antennas,as shown in FIGS. 11A-B, TX output from j^(th) element can berepresented as:

$\begin{matrix}{{X_{j}(t)} = {\sum\limits_{k = 1}^{N_{s}}{{x_{k}(t)}A_{k,j}e^{j({{\omega_{RF}t} + \psi_{k,j}})}}}} & (9)\end{matrix}$where:ω_(RF) is the operating frequency; andA_(k) and ψ_(k) are gain and phase shifts applied to k^(th) stream.

For STAR system, the aggregated TX leakage at i^(th) RX antenna can berepresented as:

$\begin{matrix}{{I_{i}(t)} = {\sum\limits_{j = 1}^{N}\left\lbrack {{❘L_{ij}❘}{e^{- {j\psi}_{ij}}\left( {\sum\limits_{k = 1}^{N_{s}}{{x_{k}\left( {t - \Delta_{ij}} \right)}e^{j({{\omega_{RF}t} + \psi_{k,j}})}}} \right)}} \right\rbrack}} & (10)\end{matrix}$

Now, to cancel the TX leakage perfectly, it can be shown that full MIMOcancellation is required (see FIG. 11A) with one independent cancelerfrom each TX element to each RX element C_(ij) as the following:

$\begin{matrix}{{I_{i}(t)} = {\left. {\sum\limits_{j = 1}^{N}\left\lbrack {C_{ij}*{X_{j}(t)}} \right\rbrack}\Rightarrow C_{ij} \right. = {{❘L_{ij}❘}{e^{- \psi_{ij}}\left( {{\delta\left( {t - \Delta_{ij}} \right)}e^{j\omega_{RF}\Delta_{ij}}} \right)}}}} & (11)\end{matrix}$

The cancellation scheme shown in the above equation and in FIG. 11A hasa number of cancellation coefficient scales with N² that makes itimpossible to realize for massive antenna array systems. Moreover, thetraditional way of performing RF cancellation from TX output to RX inputis also not possible to implement for massive antenna arrays because ofextremely complex RF routing requirement.

However, in case of phased array systems where TX and RX antennas areclosely packed with λ/2 spacing, the group delay variation of the TXleakage across antenna array is minimal and Δ_(ij)(d_(ij), θ_(ij)) fori^(th) RX and j^(th) TX antenna pair can be approximated as Δ_(o)(d_(A), θ_(A)). Therefore, aggregated leakage of Eq. (10) can beapproximated as:

$\begin{matrix}{{{I_{i}(t)} \cong {\sum\limits_{k = 1}^{N_{S}}\left\lbrack {{x_{k}\left( {t - \Delta_{o}} \right)}e^{j\omega_{RF}t}{\sum\limits_{j = 1}^{N}\left( {{❘L_{ij}❘}e^{{- j}\psi_{ij}} \times A_{k,j}e^{j\psi_{k,j}}} \right)}} \right\rbrack}} = {{{\sum\limits_{k = 1}^{N_{s}}\left\lbrack {C_{k,i}*{x_{k}(t)}\left( e^{j\omega_{RF}t} \right.} \right\rbrack}C_{k,i}} = {{\delta\left( {t - \Delta_{o}} \right)}{\sum\limits_{j = 1}^{N}{{❘L_{ij}❘}A_{k,j}e^{j({\psi_{k,j} - \psi_{ij} + {\omega_{RF}\Delta_{o}}})}}}}}} & (12)\end{matrix}$

The above equation shows that phased array system cancellation can beperformed by injecting on independent cancellation term from each TXstreams to each RX antenna, as shown in FIG. 11A. Therefore, thecancellation complexity scales with O(N) and no direct RX feeding fromTX output to RX input is required. Herein is disclosed a circuitarchitecture that intelligently realizes cancellation scheme in FIG. 11Bwith low hardware overhead. The current implementation is limited to onestream and can be scaled up flowing the system concept shown in FIG.11B.

Full-Duplex Beamforming with Three-Step Self Interference Cancellation

The STAR beamforming system shown in FIG. 12 includes separate TXantenna array (TX path is shown in blue), separate RX antenna array (RXpath is shown in red), and a built-in per-element SIC mechanism (SICpath is shown in green). The FC-HBF architecture enables the built-inSIC mechanism to support a STAR operation where half of the array isconfigured in single-stream TX mode and the other half is configured insingle-stream RX mode. The second available stream in the RX array isconfigured in the TX mode to inject a copy of the TX signal afterindependent complex weighting in each path to perform per-element SIC.Such SIC, and hence such STAR operation, is only available in an FC-HBFand not in PC-HBF. The multi-layer HBF architecture previously discussedherein, also supports such STAR operation if FC tiles are used in thefirst stage. Moreover, as shown in FIG. 12 , in a multi-layerarchitecture, because multiple baseband RX outputs are available, MIMOoperation can be supported in the RX array along with STAR operation.Please note that, although the TX array has multiple available streamsto performs MIMO, as only one independent SIC stream is available in theRX array for two-stream FC tiles, cancellations for both TX streamswould not be possible. Hence MIMO in the TX path can only be performedalong with SIC for both streams if FC-tile with more than two-streamsare used.

Disclosed herein is a successive SIC mechanism is introduced for theSTAR beamforming system, as shown in FIG. 15 , that performs RF-domaincancellation in three steps.

Step #1—Per-Element SIC

The first SIC step is most important since it cancels the leakageearliest in the RX path. As shown in FIG. 13 , in the first SIC step,one stream in the RX beamformer is used to inject complex-weighted TXsignals independently at the output node of each LNA. Assume that thebaseband equivalent representation of complex signals at the n-th LNAoutput is d_(n) and at the per-element SIC input is x_(T). The signald_(n) has two parts: the desired receive signal r_(n) and the TX leakageinterference i_(n). Now, in the first step, the SIC weights C₁-C_(N)should be set to reduce SI for each element as below.

$\begin{matrix}{{\min\limits_{C_{n}}{{{i_{n} - {C_{n}x_{T}}}}^{2}\mspace{14mu}{for}\mspace{14mu}{all}\mspace{14mu} n}} \in \left( {1,\ N} \right)} & (13)\end{matrix}$

Because the SIC signal in the first step is approximated using only asingle un-delayed complex tap, this step can only cancel SI with smallgroup delay.

Step #2—Null-Steering SIC

In the second SIC step, the beamforming degrees of freedom in the RXarray are used to minimize the residual SI after beamforming. Hence, thebeamforming weights are set so that the desired signal SNR is maximizedand the residual SI is minimized, thereby improving overallsignal-to-interference-ratio. For desired RX output symbols s_(R),second step SIC weights R₁-R_(N) should be set based on the Eq. (14):

$\begin{matrix}{{\min\limits_{\lbrack{R_{1},\ldots,R_{N}}\rbrack}{❘{S_{R} - {\sum\limits_{n = 1}^{N}{R_{N}\left( {d_{n} - {C_{n}x_{T}}} \right.}}}❘}^{2}{where}d_{n}} = {r_{n} + i_{n}}} & \end{matrix}$

-   -   (14)

In the null-steering operation, SI signals that have similar groupdelays are used to cancel each other by beamforming. Therefore, thesecond cancellation step can cancel for wider signal bandwidth. Pleasenote that the on-chip RF-domain adaptive null/beam steering technique,can be followed here to perform autonomous adaptation of second step SICweights based on Eq. (14).

Step #3—RF-SIC after Beamforming

In the third SIC step, another SIC signal is injected after theRF-domain RX beamforming at the input of the downconversion stage. ForTX copy x_(T1), the third step SIC weights C_(B) should be set based onthe Eq. (15):

$\begin{matrix}{{\min\limits_{C_{B}}\left| {S_{R} - \left\{ {\left( {\sum\limits_{n = 1}^{N}{R_{n}\left( {d_{n} - {C_{n}x_{T}}} \right)}} \right) - {C_{B}x_{T1}}} \right\}} \right.}❘}^{2} & (15)\end{matrix}$

As the thirst step SIC weight is implemented in analog or digitalbaseband, a multi-tap filter can be used (including delayed tap) in theSIC path, thereby enabling cancellation for SI leakage even with largegroup delay. The cancellation weight C_(B) can also be adapted using LMScriterion.

Note that all aforementioned SIC's are performed in the RF domain, andadditional SIC can be performed in the analog and digital domain.

Full-Duplex Beamforming Circuit Architecture

A 28/37 GHz fully connected hybrid beamforming MIMO transceiverprototype was developed to demonstrate full-duplex operation withthree-step self-interference cancellation. The prototype consists ofeight elements that are segmented into two tiles, each with four-elementtwo-stream fully connected beamforming. The prototype uses a fullybidirectional signal path where each element can be configured as eitherTX or RX. Each stream for each element (every element is connected totwo beamforming streams) can be configured either to transmit or toreceive. This extremely flexible circuit architecture, shown in FIG. 14, allows two-stream half-duplex mode operation as well as can bereconfigured to full-duplex operation with built-in three-step SICwithout any additional hardware. Therefore, the same circuit can bereconfigured to one of the two system modes (help duplex or full duplex)based on the link SNR. The circuit architecture uses homodyne Cartesiansplitting/combining based hybrid beamforming for compactness anddual-frequency operation. The circuit schematic for only a four-elementtile is shown in FIG. 14 where two elements are configured as TX (shownin blue), two elements as RX (shown in red), and the second stream inthe RX array as SIC (shown in green). The unused blocked are powereddown and shown in gray.

First Step SIC

First step SIC weight has two parts: (1) coarse grain Cartesian weight;and (2) fine-grain polar weight. The Cartesian weight is implementedbased on the Cartesian splitting beamforming approach with 5-bit(including sign bit) resolution for both the I and Q path. To get muchfiner complex weighting resolution, addition polar RF weights are usedthat consists of one 5-bit (excluding sign) addition programmable gainamplifier for gain control and 5-bit phase control. The fine phasecontrol is implemented using a digitally controllable capacitor bank inthe coupled resonator tank, as shown in FIG. 14 , and can perform phasetuning with <0.5° resolution and ˜15° dynamic range. The first step SICcurrent signal is injected in the secondary port of the coupledresonator load whose primary port is connected to the forward path LNAfinal stage.

Second Step SIC

The second SIC step is based on beamforming in the RX array. Thebeamforming weights are implemented using the Cartesian combing approachwith the same weighting principle as the first step SIC, i.e., 5-bit I/QCartesian weights, and additional 5-bit fine gain and 5-bit phase weightconfigured in the receive mode.

Third Step SIC

The third SIC step is implemented using baseband Cartesian complexweight with 6-bit resolution in both the I and Q path. The up- anddown-conversion Cartesian based complex mixing stages have independenthardware for bidirectional operation, and only one of them is turned onfor TX or RX operation. Hence, the up-conversion mixer hardware isalready available in RX stream, which is repurposed as third step SIC inthis circuit architecture without requiring any extra hardware. The SICcurrent is injected at the input of the mixer at one port of a couplingresonator, similar to the first SIC step.

Least-Mean-Square-Based RF-Domain SIC Adaptation

The per-element SIC weight in the first step canceller can bedynamically computed under a minimum-mean-square-error criterion tominimize the residual SI using an LMS algorithm. The LMS-based SICweight update algorithm can be expressed as:

$\begin{matrix}{{C\left( {k + 1} \right)} = {{{C(k)} - {\mu{\nabla_{C}\left\lbrack {\sum\limits_{i = 1}^{N}{❘{{d_{i}(k)} - {{C_{i}(k)}{x(k)}}}❘}^{2}} \right\rbrack}}} = {{C(k)} + {2{\mu\left\lbrack {{d(k)} - {{C(k)}{x(k)}}} \right\rbrack}^{*}{x(k)}}}}} & (16)\end{matrix}$where:μ is the adaptation rate;C(k) is the cancellation weight;x(k) is the transmitted symbol; andthe d(k) vector is the leakage input to the RX array at the kth timeinstance.

Eq. (16) indicates that, to compute the updated weight of each RXelement's canceller, access to the error signal (i.e., the residual SIafter the per-element cancellation) for each receive element isrequired. However, because the received signals from all RX elements arecombined before downconversion and digitization in an FC-HBF-based STARbeamforming system (as in FIG. 13 ), the baseband equivalent errorsignal for each RX element is not available to the SIC adaptationengine. Here we propose a time-multiplexed error-extraction scheme thatbreaks this requirement.

Time-Multiplexed Error Extraction

The time-multiplexed error-extraction scheme shares a similar underlyingconcept with the time-multiplexed LMS beam adaption. In thetime-multiplexed SIC scheme, each row of the vector equation (Eq. (16))that corresponds to one antenna path can be computed at a separate timeinstance sequentially. The error signal for each antenna path can,therefore, be sequentially extracted by setting the RX beamformingweight of the corresponding path to unity and the other paths to 0, asshown in FIG. 15 . Therefore, in the time-multiplexed SIC scheme, onefull adaptation cycle requires N-baseband cycles for N RX elements(shown in FIG. 15 for four RX antennas). In this design, thesign-sign-LMS scheme is implemented for simplicity, where a pair of I/Qcomparators are used to extract the TX signal and the RX error (shown inFIG. 16A for single element]. The comparator and the digital SIC engineare strobed at opposite clock edge to provide half the time period ofthe baseband clock to the digital and the analog portions to settle. Ashortfall to the time-multiplexed SIC scheme is that the adaptation timeincreases with the number of RX elements. This can be improved by usingour proposed scalable, tiled HBF architecture with FC-tile, where theadaptation rate can be improved by K-times for K RX tiles.

One-Step Non-Ideality Correction

FIG. 16B shows the baseband equivalent representation of the per-elementSIC adaptation loop for a single element scenario. Here, complex-weightsA and B model the gain and phase shifts of the signal's complex envelopin the TX and RX paths, respectively. Eq. (16) can now be rewrittenincluding A and B for a single element as follows:C(k+1)=C(k)−μ∇_(C)[|B(d(k)−C(k)x(k)A)|²]=C(k)+2μ[e(k)*][x(k)AB]   (17)

Although any gain component from the correction factor AB to the TXoutput x(k) doesn't require calibration, any phase component in thispath, if not appropriately accounted, can cause incorrect operation andmake the LMS loop not to converge. The correction factor AB can beestimated by transmitting a known symbol (e.g., x) from the transmitterwith a unity setting of C's and measuring the output symbol from the RXpath (i.e., xAB). Therefore, in the case of sign-sign-LMS, sign(xAB)needs to be estimated once for every possible transmitted symbol (e.g.,the calibration requires four cycles for SIC adaptation with QPSKsymbols). Because the correction factor AB is approximately the same foreach element in the RX array due to symmetry and matching, thecorrection factor needs to be estimated only once before the beamadaptation cycle in FIG. 15 , which makes it low calibration overhead.

Fine-Resolution Polar Weight Update

The disclosed invention prototype uses 5-bit I-path and 5-bit Q-pathCartesian weight (weight implementation is described in Section IV andV). The SIC weight update algorithm in Eq. (17) that is already in theCartesian form can therefore be directly applied to the Cartesian SICweights without any additional digital computation. However, to performfiner gain and phase control, the prototype also incorporates additionalhigh-resolution polar gain and phase controls. The fine-grain polarweight (A, θ) set the overall SIC path complex weight only around theproximity of the coarse-grain Cartesian weight (C_(I), C_(Q)) with highresolution. The polar weight updates (ΔA, Δθ) can be calculated from theknown Cartesian weight updates (ΔC_(I), ΔC_(Q)) by solving the Eq. (18):(C _(I) +ΔC _(I))+j(C _(Q) +ΔC _(Q))=(A+ΔA)e ^(j(θ+θΔ))   (18)where:

${{A = \sqrt{C_{Q}^{1} + C_{Q}^{2}}};{and}}{\theta = {\tan^{- 1}\left( \frac{C_{Q}}{C_{I}} \right)}}$

The solution to the above equation is the following:

$\begin{matrix}{{{\Delta A} = \frac{\left( {{C_{I} \times {\Delta C}_{I}} + {C_{Q} \times \Delta C_{Q}}} \right)}{A}}{{\Delta\theta} = \frac{\left( {{C_{I} \times \Delta C_{Q}} - {C_{Q} \times {\Delta C}_{I}}} \right)}{A^{2}}}} & (19)\end{matrix}$

To reduce the computational complexity, only the sign of the polarweight update can be used. Hence, the gain can be updated asA+μ_(Ax)sign(ΔA), and the phase can be updated as θ+μ_(θx)sign(Δθ),where μ_(A) and μ_(θ) are adaptation rates. Note that the C_(I) andC_(Q) values are not updated during multiple cycles of polar weightupdate, because A and θ are adapted only across the vicinity of C₁ andC_(Q) within one LSB resolution. Additionally, note that the polarweight update is also performed for each element sequentially after thecoarse Cartesian weight is settled.

Self-Interference Cancellation Using Group Delay Correction

A potential solution to put an equivalent delay in the digital domain toimprove the SIC performance across signal bandwidth, even in thepresence of these group delays from various components, will now bedisclosed. The first step SIC mechanism is shown in FIG. 17 along withall group delay components. This mechanism may also be used in the thirdstep. After complex weighting in the TX beamformer, the TX path signaltravels through various RF elements and acquires group delays as thefollowing: D_(PA) from the PA, D_(CON) from the connector from the PAoutput to the TX antenna, D_(LEAK) from the TX array to the RX array,D_(CON) from the connector from the RX antenna to the RX input, andfinally, D_(LNA) from the LNA. Therefore, with respect to the SIC copyin the RX array, the SI signal has additional group delay ofD_(TOT)=D_(PA)+2D_(CON)+D_(LEAK)+D_(LNA).

In the case of an FD system with tiled multi-layer architecture (similarto that shown in FIG. 12 ), if separate tiles are used for the TX andthe RX array, the TX path and the SIC path will have separateupconverter, as shown in FIG. 17 . As such, one can put a delay in thedigital domain (say, D_(TX)) and feed a delayed version of the TX copyto the RX array for SIC. Because D_(PA), D_(CON), and D_(LNA) are fixedfor any pair of TX and RX antenna paths, a single fixed D_(TX)(=D_(PA)+2D_(CON)+D_(LNA)) in digital domain can compensate these groupdelays at once for all RX elements. Assuming digital delay resolution of0.4 ns (for 250 Msym/s data rate and 10 over-sampling-ratio), worst caseresidual group delay can be approximated as 0.4 ns. Therefore, even whensignificant group delays are present, proposed digital domain delaycompensation technique can help to achieve over 20 dB SIC for 300 MHz RFBW.

Disclosed herein is a multi-layer MIMO/beamforming architecture thatfacilitates efficient scaling in the number of antennas and streamswithout degrading system performance and complexity comprising amulti-layer architecture that employs a fully-connected (FC) “tile” thatsimultaneously enables inter-band carrier aggregation (dual-bandoperation) and multi-stream MIMO in each band. The multi-layerarchitecture employs an FC-tile that enables full-duplex communicationwith MIMO capability in the receive path, phased-array beamforming(single-stream) in the transmit path, per-antenna self-interferencecancellation (SIC) in each RX element of TX signal-leakage andper-stream self-interference cancellation (SIC) in each RXdownconversion chain of TX signal-leakage. In addition, a three step SICtechnique to perform successive SIC in multi-antenna full-duplex systemis disclosed. Lastly, an autonomous SIC weight update algorithm forper-element SIC step has been disclosed.

To those skilled in the art to which the invention relates, manymodifications and adaptations of the invention will suggest themselves.Implementations provided herein, including sizes, shapes, ratings andspecifications of various components or arrangements of components, anddescriptions of specific manufacturing processes, should be consideredexemplary only and are not meant to limit the invention in any way. Asone of skill in the art would realize, many variations onimplementations discussed herein which fall within the scope of theinvention are possible.

The invention claimed is:
 1. A multi-layer, MIMO beamforming transceivercomprising: K tiles comprising N_(A) antenna elements; a first layerinterfacing the N_(A) antenna elements to N_(RF) frequency translationchains; and a second layer implementing fully-connected, bi-directionalspatial signal processing between the N_(RF) frequency translationchains and N_(S) streams; wherein the K tiles are fully-connected; andwherein N_(RF)=2K.
 2. The transceiver of claim 1 wherein the first layerapplies RF-domain complex weighting to the N_(RF) frequency translationchains.
 3. The transceiver of claim 1 wherein the spatial signalprocessing in the second layer occurs in the analog domain.
 4. Thetransceiver of claim 1 wherein the fully-connected beamforming in thesecond layer occurs in the digital domain.
 5. The transceiver of claim 1wherein complex weights are applied to received frequency translationchains to spatially separate the NS streams.
 6. The transceiver of claim1 comprising a receive path using a Cartesian combining architecture toperform RF domain beamforming in each stream of each fully-connectedtile in the first layer.
 7. The transceiver of claim 1 comprising atransmit path using a Cartesian-splitting architecture to perform RFdomain beamforming in each stream of each fully-connected tile in thefirst layer.
 8. The transceiver of claim 1 further comprising: alow-noise amplifier/power amplifier dual-band bi-directional interfaceconnected to each antenna element.
 9. The transceiver of claim 8 furthercomprising: a dual-band bi-directional beamforming network with sharedpassives between transmit and receive paths, coupled to the low-noiseamplifier/power amplifier interfaces.
 10. The transceiver of claim 9further comprising: N_(S) homodyne complex-quadrature up/downconversions stages, coupled to the beamforming network, one stage perstream respectively.
 11. The transceiver of claim 10 further comprising:a dual-band local oscillation generation and distribution network. 12.The transceiver of claim 11 further comprising: a three-stepself-interference cancellation (SIC) mechanism.
 13. The transceiver ofclaim 12 wherein a first step of the three-step SIC mechanism comprises:using one stream in each receive beamformer to inject a complex-weightedtransmit signal at an output node of each low-noise amplifier.
 14. Thetransceiver of claim 13 wherein the complex weights are dynamicallycomputed using a minimum-mean-square-error criterion to minimize theresidual self-interference using a least-mean-squared algorithm.
 15. Thetransceiver of claim 14 wherein the least-mean-squared algorithm usestime-multiplexed error extraction.
 16. The transceiver of claim 13wherein a second step of the three-step SIC mechanism comprises: usingbeamforming degrees of freedom in the receive portion of the beamformingnetwork to minimize residual self-interference after beamforming. 17.The transceiver of claim 16 wherein a third step of the three-step SICmechanism comprises: injecting a self-interference cancellation signalafter the RF-domain receive beamforming at an input of a downconversionstage.
 18. The transceiver of claim 17 wherein further self-interferencecancellation is performed at baseband stage in either the analog ordigital domains.
 19. The transceiver of claim 17 wherein the first andthird step cancellations use a transmit signal copy with digital domaindelay to achieve wideband self-interference cancellation.
 20. Amulti-layer, MIMO beamforming transceiver comprising: K tiles comprisingN_(A) antenna elements; a first partially-connected layer interfacingthe N_(A) antenna elements to N_(RF) frequency translation chains; and asecond fully-connected analog layer implementing bi-directional spatialsignal processing between the N_(RF) frequency translation chains andN_(S) streams; wherein N_(RF)=K.
 21. A multi-layer, MIMO beamformingtransceiver comprising: K tiles comprising N_(A) antenna elements; afirst fully-connected layer interfacing the N_(A) antenna elements toN_(RF) frequency translation chains; and a second fully-connecteddigital layer implementing bi-directional spatial signal processingbetween the N_(RF) frequency translation chains and the N_(S) streams;wherein N_(RF)=2K.
 22. A multi-layer, MIMO beamforming transceivercomprising: K tiles comprising N_(A) antenna elements; a firstfully-connected layer interfacing the N_(A) antenna elements to N_(RF)frequency translation chains; and a second fully-connected analog layerimplementing bi-directional spatial signal processing between the N_(RF)frequency translation chains and the N_(S) streams; wherein N_(RF)=2K.23. The transceiver of claim 22 further comprising: a third,fully-connected digital layer coupled to the second layer via aplurality of analog-to-digital converters.